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The discrete Fourier transform (DFT) is one of the two most common, and powerful, procedures encountered in the field of digital signal processing. (Digital filtering is the other.) The DFT enables us to analyze, manipulate, and synthesize signals in ways not possible with continuous (analog) signal processing. Even though it's now used in almost every field of engineering, we'll see applications for DFT continue to flourish as its utility becomes more widely understood. Because of this, a solid understanding of the DFT is mandatory for anyone working in the field of digital signal processing. The DFT is a mathematical procedure used to determine the harmonic, or frequency, content of a discrete signal sequence. Although, for our purposes, a discrete signal sequence is a set of values obtained by periodic sampling of a continuous signal in the time domain, we'll find that the DFT is useful in analyzing any discrete sequence regardless of what that sequence actually represents. The DFT's origin, of course, is the continuous Fourier transform X(f) defined as (eqn. 1) where x(t) is some continuous time-domain signal.± In the field of continuous signal processing, (eqn. 1) is used to trans form an expression of a continuous time-domain function x(t) into a continuous frequency-domain function X(f). Subsequent evaluation of the X(f) expression enables us to determine the frequency content of any practical signal of interest and opens up a wide array of signal analysis and processing possibilities in the fields of engineering and physics. One could argue that the Fourier transform is the most dominant and widespread mathematical mechanism available for the analysis of physical systems. (A prominent quote from Lord Kelvin better states this sentiment: "Fourier's theorem is not only one of the most beautiful results of modern analysis, but it may be said to furnish an indispensable instrument in the treatment of nearly every recondite question in modern physics." By the way, the history of Fourier's original work in harmonic analysis, relating to the problem of heat conduction, is fascinating. References [1] and [2] are good places to start for those interested in the subject.) With the advent of the digital computer, the efforts of early digital processing pioneers led to the development of the DFT defined as the discrete frequency-domain sequence X(m), where DFT equation (exponential form): (eqn. 2) For our discussion of (eqn. 2), x(n) is a discrete sequence of time-domain sampled values of the continuous variable x(t). The "e" in (eqn. 2) is, of course, the base of natural logarithms and j = √ -1. 1 Equation (eqn. 2) has a tangled, almost unfriendly, look about it. Not to worry. After studying this section, (eqn. 2) will become one of our most familiar and powerful tools in understanding digital signal processing. Let's get started by expressing (eqn. 2) in a different way and examining it carefully. From Euler's relationship e^-j theta = cos(theta) -jsin(theta), (eqn. 2) is equivalent to (eqn. 3) We have separated the complex exponential of (eqn. 2) into its real and imaginary components where X(m) = the mth DFT output component, i.e., X(0), X(1), X(2), X(3), etc., m = the index of the DFT output in the frequency domain, m = 0, 1, 2, 3, . ., N-1, x(n) = the sequence of input samples, x(0), x(1), x(2), x(3), etc., n = the time-domain index of the input samples, n = 0, 1, 2, 3, . N-1, j = √-1 , and N = the number of samples of the input sequence and the number of frequency points in the DFT output. Although it looks more complicated than (eqn. 2), (eqn. -3) turns out to be easier to understand. (If you're not too comfortable with it, don't let the j = √-1 i concept bother you too much. It's merely a convenient abstraction that helps us compare the phase relationship between various sinusoidal components of a signal. Section 8 discusses the j operator in some detail.)t The indices for the input samples (n) and the DFT output samples (m) always go from 0 to N-1 in the standard DFT notation. This means that with N input time-domain sample values, the DFT determines the spectral content of the input at N equally spaced frequency points. The value N is an important parameter because it determines how many input samples are needed, the resolution of the frequency-domain results, and the amount of processing time necessary to calculate an N-point DFT. It's useful to see the structure of (eqn. 3) by eliminating the summation and writing out all the terms. For example, when N = 4, n and m both go from 0 to 3, and (eqn. 3) becomes (eqn. 4a) Writing out all the terms for the first DFT output term corresponding to m=0, (eqn. 4b) For the second DFT output term corresponding to m = 1, (eqn. 4a) becomes (eqn. 4c) Instead of the letter j, be aware that mathematicians often use the letter i to represent √-1 the operator. For the third output term corresponding to m = 2, (eqn. 4a) becomes (eqn. 4d) Finally, for the fourth and last output term corresponding to m = 3, Eq. eqn. 4a) becomes (eqn. 4e) The above multiplication symbol "." in (eqn. 4) is used merely to separate the factors in the sine and cosine terms. The pattern in (eqn. 4b) through (eqn. 4e) is apparent now, and we can certainly see why it's convenient to use the summation sign in (eqn. 3). Each X(m) DFT output term is the sum of the point for point product between an input sequence of signal values and a complex sinusoid of the form cos(0) - jsin(0). The exact frequencies of the different sinusoids depend on both the sampling rate f s, at which the original signal was sampled, and the number of samples N. For example, if we are sampling a continuous signal at a rate of 500 samples/s and, then, perform a 16-point DFT on the sampled data, the fundamental frequency of the sinusoids is fs /N = 500/16 or 31.25 Hz. The other X(m) analysis frequencies are integral multiples of the fundamental frequency, i.e., X(0) = 1st frequency term, with analysis frequency = 0 • 31.25 = 0 Hz, X(1) = 2nd frequency term, with analysis frequency = 1 • 31.25 = 31.25 Hz, X(2) = 3rd frequency term, with analysis frequency = 2 • 31.25 = 62.5 Hz, X(3) = 4th frequency term, with analysis frequency =3 • 31.25 = 93.75 Hz, X(15) = 16th frequency term, with analysis frequency = 15 • 31.25 = 468.75 Hz. The N separate DFT analysis frequencies are mfs (eqn. 5)
So, in this example, the X(0) DFT term tells us the magnitude of any 0-Hz ("DC") component contained in the input signal, the X(1) term specifies the magnitude of any 31.25-Hz component in the input signal, and the X(2) term indicates the magnitude of any 62.5-Hz component in the input signal, etc. Moreover, as we'll soon show by example, the DFT output terms also deter mine the phase relationship between the various analysis frequencies contained in an input signal. Quite often we're interested in both the magnitude and the power (magnitude squared) contained in each X(m) term, and the standard definitions for right triangles apply here as depicted in FIG. 1. If we represent an arbitrary DFT output value, X(m), by its real and imaginary parts (eqn. 6) the magnitude of X(m) is
By definition, the phase angle of X(m), Xo(m), is I X. ' (eqn. 8) The power of X(m), referred to as the power spectrum, is the magnitude squared where (eqn. 9) 1.1 The above Eqs. (eqn. 2) and (eqn. 3) will become more meaningful by way of an ex- ample, so, let's go through a simple one step-by-step. Let's say we want to sample and perform an 8-point DFT on a continuous input signal containing components at 1 kHz and 2 kHz, expressed as (eqn. 10) To make our example input signal xi (t) a little more interesting, we have the 2-kHz term shifted in phase by 135° (37c/4 radians) relative to the 1-kHz sinewave. With a sample rate of fs, we sample the input every 1/4 = t, seconds. Because N = 8, we need 8 input sample values on which to perform the DFT. So the 8-element sequence x(n) is equal to x 1 (t) sampled at the nts instants in time so that (eqn. 11) If we choose to sample x1 (t) at a rate of f5 = 8000 samples /s from (eqn. 5), our DFT results will indicate what signal amplitude exists in x(n) at the analysis frequencies of mfs /N, or 0 kHz, 1 kHz, 2 kHz,. . ., 7 kHz. With f, = 8000 samples/s, our eight x(n) samples are (eqn. 11') These x(n) sample values are the dots plotted on the solid continuous xi (t) curve in FIG. 2(a). (Note that the sum of the sinusoidal terms in (eqn. 10), shown as the dashed curves in FIG. 2(a), is equal to xin(t).) Now we're ready to apply (eqn. 3) to determine the DFT of our x(n) input. We'll start with m = 1 because the m = 0 case leads to a special result that we'll discuss shortly. So, for m = 1, or the 1-kHz (rnfs /N = 1.800018) DFT frequency term, (eqn. 3) for this example becomes (eqn. 12) Next we multiply x(n) by successive points on the cosine and sine curves of the first analysis frequency that have a single cycle over our 8 input samples. In our example, for m = 1, we'll sum the products of the x(n) sequence with a 1-kHz cosine wave and a 1-kHz sinewave evaluated at the angular values of 2nn /8. Those analysis sinusoids are shown as the dashed curves in FIG. 2(b). Notice how the cosine and sinewaves have m = 1 complete cycles in our sample interval. Substituting our x(n) sample values into (eqn. 12) and listing the cosine terms in the left column and the sine terms in the right column, we have ... So we now see that the input x(n) contains a signal component at a frequency of 1 kHz. Using (eqn. 7), (eqn. 8), and (eqn. 9) for our X(1) result, Xmag(1) = 4, Xps(1) = 16, and X(1)'s phase angle relative to a 1-kHz cosine is X0(1) = -90°. For the m = 2 frequency term, we correlate x(n) with a 2-kHz cosine wave and a 2-kHz sinewave. These waves are the dashed curves in FIG. 2(c). Notice here that the cosine and sinewaves have m = 2 complete cycles in our sample interval in FIG. 2(c). Substituting our x(n) sample values in (eqn. 3), for m = 2, gives ... Here our input x(n) contains a signal at a frequency of 2 kHz whose relative amplitude is 2, and whose phase angle relative to a 2-kHz cosine is 45°. For the 111 = 3 frequency term, we correlate x(n) with a 3-kHz cosine wave and a 3-kHz sinewave. These waves are the dashed curves in FIG. 2(d). Again, see how the cosine and sinewaves have in = 3 complete cycles in our sample interval in FIG. 2(d). Substituting our x(n) sample values in (eqn. 3) for in = 3 gives ... Our DFT indicates that x(n) contained no signal at a frequency of 3 kHz. Let's continue our DFT for the m = 4 frequency term using the sinusoids in FIG. 3(a). So (eqn. 3) is FIG. 3 DFT Example 1: (a) the input signal and the m = 4 sinusoids; (b) the Input and the m =5 sinusoids; (c) the input and the m = 6 sinusoids; (d) the Input and the m = 7 sinusoids. Our DFT for the m = 5 frequency term using the sinusoids in FIG. 3(h) yields .... For the m =6 frequency term using the sinusoids in FIG. 3(c), (eqn. 3) is ... If we plot the X(m) output magnitudes as a function of frequency we get the magnitude spectrum of the x(n) input sequence, shown in FIG. 4(a). The phase angles of the X(m) output terms are depicted in FIG. 4(b). Hang in there, we're almost finished with our example. We've saved the calculation of the m = 0 frequency term to the end because it has a special significance. When m = 0, we correlate x(n) with cos(0) - jsin(0) so that (eqn. 3) becomes (eqn. 13) Because cos(0) = 1, and sin(0) = 0, (eqn. 13') We can see that (eqn. 13') is the sum of the x(n) samples. This sum is, of course, proportional to the average of x(n). (Specifically, X(0) is equal to N times x(n)'s average value). This makes sense because the X(0) frequency term is the nontime-varying (DC) component of x(n). If X(0) were nonzero, this would tell us that the x(n) sequence is riding on a DC bias and has some nonzero average value. For our specific example input from (eqn. 10), the sum, however, is zero. The input sequence has no DC component, so we know that X(0) will be zero. But let's not be lazy-we'll calculate X(0) anyway just to be sure. Evaluating (eqn. 3) or (eqn. 13') for rn = 0, we see that ... So our x(n) had no DC component, and, thus, its average value is zero. Notice that FIG. 4 indicates that xin(t), from (eqn. 10), has signal components at 1 kHz (m = 1) and 2 kHz (m = 2). Moreover, the 1-kHz tone has a magnitude twice that of the 2-kHz tone. The DFT results depicted in FIG. 4 tell us exactly the spectral content of the signal defined by Eqs. (eqn. 10) and (eqn. 11). The perceptive reader should be asking two questions at this point. First, what do those nonzero magnitude values at m = 6 and m = 7 in FIG. 4(a) mean? Also, why do the magnitudes seem four times larger than we would expect? We'll answer those good questions shortly. The above 8-point DFT ex ample, although admittedly simple, illustrates two very important characteristics of the DFT that we should never forget. First, any individual X(m) output value is nothing more than the sum of the term-by-term products, a correlation, of an input signal sample sequence with a cosine and a sinewave whose frequencies are m complete cycles in the total sample interval of N samples. This is true no matter what the f, sample rate is and no matter how large N is in an N-point DFT. The second important characteristic of the DFT of real input samples is the symmetry of the DFT output terms. 2. Looking at FIG. 4(a) again, there is an obvious symmetry in the DFT results. Although the standard DFT is designed to accept complex input sequences, most physical DFT inputs (such as digitized values of some continuous signal) are referred to as real, that is, real inputs have nonzero real sample values, and the imaginary sample values are assumed to be zero. When the input sequence x(n) is real, as it will be for all of our examples, the complex DFT outputs for m = 1 to m = (N/2) - 1 are redundant with frequency output values for m> (N/2). The mth DFT output will have the same magnitude as the (N-m)th DFT output. The phase angle of the DFT's mth out- put is the negative of the phase angle of the (N-m)th DF .1' output. So the mth and (N-m)th outputs are related by the following: X(m) =I X(m) I at X,,(m) degrees (eqn. 14) We can state that when the DFT input sequence is real, X(m) is the complex conjugate of X(N-m), or X(m)= X * (N - m) (eqn. 14) where the superscript symbol denotes conjugation. In our example above, notice in Figures 3-4(b) and 3-4(d) that X(5), X(6), and X(7) are the complex conjugates of X(3), X(2), and X(1), respectively. Like the DFT's magnitude symmetry, the real part of X(m) has what is called even symmetry, as shown in FIG. 4(c), while the DFT's imaginary part has odd symmetry, as shown in FIG. 4(d). This relationship is what is meant when the DFT is called conjugate symmetric in the literature. It means that, if we perform an N-point DFT on a real input sequence, we'll get N separate complex DFT output terms, but only the first N/241 terms are independent. So to obtain the DFT of x(n), we need only compute the first N/2+1 values of X(m) where 0 nt (N/2); the X(N/2+1) to X(N-1) DFT output terms provide no additional information about the spectrum of the real sequence x(n). [Using our notation, the complex conjugate of x = a + jb is defined as x. = a - jb; that is, we merely change the sign of the imaginary part of x. In an equivalent form, if x = ei", then x. = e ] Although Eqs. (eqn. 2) and (1-3) are equivalent, expressing the DFT in the exponential form of (eqn. 2) has a terrific advantage over the form of (eqn. 3). Not only does (eqn. 2) save pen and paper, (eqn. 2)'s exponentials are much easier to manipulate when we're trying to analyze DFT relationships. Using (eqn. 2), products of terms become the addition of exponents and, with due respect to Euler, we don't have all those trigonometric relationships to memorize. Let's demonstrate this by proving (eqn. 14) to show the symmetry of the DFT of real input sequences. Substituting N-m for m in (eqn. 2), we get the expression for the (N-m)th component of the DFT: Because CP" = cos(27cn) -jsin(27tn) = 1 for all integer values of n, (eqn. 15') We see that X(N-m) in (eqn. 15') is merely X(m) in (eqn. 2) with the sign re versed on X(m)'s exponent-and that's the definition of the complex conjugate. This is illustrated by the DFT output phase-angle plot in FIG. 4(b) for our DFT Example 1. Try deriving (eqn. 15') using the cosines and sines of (eqn. 3), and you'll see why the exponential form of the DFT is so convenient for analytical purposes. There's an additional symmetry property of the DFT that deserves mention at this point. In practice, we're occasionally required to determine the DFT of real input functions where the input index n is defined over both positive and negative values. If that real input function is even, then X(m) is al ways real and even; that is, if the real x(n) = x(-n), then, Xreal (m) is in general nonzero and Xi. mag(M) is zero. Conversely, if the real input function is odd, x(n) = -x(-n), then X 1 (m) is always zero and Ximag(m) is, in general, nonzero. This characteristic of input function symmetry is a property that the DFr shares with the continuous Fourier transform, and (don't worry) we'll cover specific examples of it later in Section 13 and in Section 5. 3. The DFT has a very important property known as linearity. This property states that the DFT of the sum of two signals is equal to the sum of the trans forms of each signal; that is, if an input sequence xi (n) has a DFT Xi (m) and another input sequence x2(n) has a DFT X2 (m), then the DFT of the sum of these sequences xsum(n) = (n) + x2 (n) is (eqn. 16) This is certainly easy enough to prove. If we plug xs .,(n) into (eqn. 2) to get X.(m), then Without this property of linearity, the DFT would be useless as an analytical tool because we could transform only those input signals that contain a single sinewave. The real-world signals that we want to analyze are much more complicated than a single sinewave. 4. The DFT Example 1 results of I X(1) I = 4 and I X(2) I = 2 may puzzle the reader because our input x(n) signal, from (eqn. 11), had peak amplitudes of 1.0 and 0.5, respectively. There's an important point to keep in mind regarding DFTs defined by (eqn. 2). When a real input signal contains a sinewave component of peak amplitude A„ with an integral number of cycles over N input samples, the output magnitude of the DFT for that particular sinewave is Mr where M r = Ao N/2 (eqn. 17) If the DFT input is a complex sinusoid of magnitude A„ (i.e., A „ei2 pi ft) with an integral number of cycles over N samples, the output magnitude of the DFT is Mc where Mc = AoN (eqn. 17') As stated in relation to (eqn. 13'), if the DFT input was riding on a DC value equal to D,9 the magnitude of the DFT's X(0) output will be DoN. Looking at the real input case for the 1000 Hz component of (eqn. 11), Ao = 1 and N = 8, so that Mra, = 1 • 8/2 = 4, as our example shows. Equation (eqn. 17) may not be so important when we're using software or floating-point hardware to perform DFTs, but if we're implementing the DFT with fixed-point hardware, we have to be aware that the output can be as large as N/2 times the peak value of the input. This means that, for real inputs, hardware memory registers must be able to hold values as large as N/2 times the maximum amplitude of the input sample values. We discuss DFT output magnitudes in further detail later in this section. The DFT magnitude expressions in Eqs. (eqn. 17) and (eqn. 17') are why we occasionally see the DFT defined in the literature as (eqn. 18) The 1/N scale factor in (eqn. 18) makes the amplitudes of X'(m) equal to half the time-domain input sinusoid's peak value at the expense of the additional division by N computation. Thus, hardware or software implementations of the DFT typically use (eqn. 2) as opposed to (eqn. 18). Of course, there are always exceptions. There are commercial software packages using (eqn. 18) for the forward and inverse DFTs. (In Section 7, we discuss the meaning and significance of the inverse DFT.) The 1 / -0 /71 scale factors in Eqs. (eqn. 18') seem a little strange, but they're used so that there's no scale change when transforming in either direction. When analyzing signal spectra in practice, we're normally more interested in the relative magnitudes rather than absolute magnitudes of the individual DFT outputs, so scaling factors aren't usually that important to us. 5. The frequency axis ni of the DFT result in FIG. 4 deserves our attention once again. Suppose we hadn't previously seen our DFT Example 1, were given the eight input sample values, from (eqn. 11'), and asked to perform an 8-point DFT on them. We'd grind through (eqn. 2) and get the X(m) values shown in FIG. 4. Next we ask, "What's the frequency of the highest magnitude component in X(m) in Hz?" The answer is not "1." The answer depends on the original sample rate f. Without prior knowledge, we have no idea over what time interval the samples were taken, so we don't know the absolute scale of the X(m) frequency axis. The correct answer to the question is to take f s and plug it into (eqn. 5) with m = 1. Thus, if fs = 8000 samples/s, then the frequency associated with the largest DFT magnitude term is If we said the sample rate f; was 75 samples/s, we'd know, from (eqn. 5), that the frequency associated with the largest magnitude term is now […] OK, enough of this-just remember that the DFT's frequency spacing (resolution) is f,/N. To 'recap what we've learned so far: • each DFT output term is the sum of the term-by-term products of an input time-domain sequence with sequences representing a sine and a cosine wave, • for real inputs, an N-point [) FT's output provides only N/2+1 independent terms, • the DFT is a linear operation, • the magnitude of the DFT results are directly proportional to N, and • the DFT's frequency resolution is fliN. It's also important to realize, from (eqn. 5), that X(N/2+ I), when ni N/2+1, corresponds to half the sample rate, i.e., the folding (Nyquist) frequency f/2. 6. There's an important property of the DFT known as the shifting theorem. It states that a shift in time of a periodic x(n) input sequence manifests itself as a constant phase shift in the angles associated with the DFT results. (We won't derive the shifting theorem equation here because its derivation is included in just about every digital signal processing textbook in print.) If we decide to sample x(n) starting at n equals some integer k, as opposed to n = 0, the DFT of those time-shifted sample values is Xshifted (m) where (eqn. 19) Equation (eqn. -19) tells us that, if the point where we start sampling x(n) is shifted to the right by k samples, the DFT output spectrum of Xshifted (m) is X(nt) with each of X(m)'s complex terms multiplied by the linear phase shift e^j2itkm/N, which is merely a phase shift of 2r.km/N radians or 360km/N degrees. Conversely, if the point where we start sampling x(n) is shifted to the left by k samples, the spectrum of Xshifted (m) is X(m) multiplied by e pain/N. Let's illustrate (eqn. 19) with an example.
6.1 Suppose we sampled our DFT Example 1 input sequence later in time by k = 3 samples. FIG. 5 shows the original input time function, x_in (t) = sin(2 pi 1000t) + 0.5sin(2 pi 200011-3 pi /4) . We can see that FIG. 5 is a continuation of FIG. 2(a). Our new x(n) sequence becomes the values represented by the solid black dots in FIG. 5 whose values are ... The values in (eqn. 21) are illustrated as the dots in FIG. 6. Notice that FIG. 6(a) is identical to FIG. 4(a). Equation (eqn. 19) told us that the magnitude of Xshifted (m) should be unchanged from that of X(m). That's a comforting thought, isn't it? We wouldn't expect the ENT magnitude of our original periodic x 1 (t) to change just because we sampled it over a different time interval. The phase of the LOT result does, however, change depending on the instant at which we started to sample xin (t). By looking at the m = 1 component of X_shifted (m), for example, we can double-check to see that phase values in Figure 6(b) are correct. Using (eqn. 19) and remembering that X(1) from DFF Example] had a magnitude of 4 at a phase angle of -90 (or -Tc/2 radians), k = 3 and N = 8 so that (eqn. 22) So X_shifted (1) has a magnitude of 4 and a phase angle of rc/4 or +45°, which is what we set out to prove using (eqn. 19). 7. Although the DFT is the major topic of this section, it's appropriate, now, to introduce the inverse discrete Fourier transform (IDFT). Typically we think of the DFT as transforming time-domain data into a frequency-domain representation. Well, we can reverse this process and obtain the original time domain signal by performing the IDFT on the X(m) frequency-domain values. The standard expressions for the IDFT are … and equally, (eqn. 23') Remember the statement we made in Section 1 that a discrete time domain signal can be considered the sum of various sinusoidal analytical frequencies and that the X(m) outputs of the DFT are a set of N complex values indicating the magnitude and phase of each analysis frequency comprising that sum. Equations (eqn. 23) and (eqn. 23') are the mathematical expressions of that statement. It's very important for the reader to understand this concept. If we perform the IDFT by plugging our results from DFT Example 1 into (eqn. 23), we'll go from the frequency-domain back to the time-domain and get our original real (eqn. 11') x(n) sample values of x(0) = 0.3535 + j0.0 x(1) = 0.3535 +j0.0 x(2) = 0.6464 + j0.0 x(3) = 1.0607 + j0.0 x(4) = 0.3535 + j0.0 x(5) = -1.0607 + j0.0 x(6) = -1.3535 +j0.0 x(7) = -0.3535 + j0.0 . Notice that (eqn. 23)'s IDFT expression differs from the DFT's (eqn. 2) only by a 1/N scale factor and a change in the sign of the exponent. Other than the magnitude of the results, every characteristic that we've covered thus far regarding the DFT also applies to the IDFT. 3.8 DFT LEAKAGE Hold on to your seat now. Here's where the DFT starts to get really interesting. The two previous DFT examples gave us correct results because the input x(n) sequences were very carefully chosen sinusoids. As it turns out, the DFT of sampled real-world signals provides frequency-domain results that can be misleading. A characteristic, known as leakage, causes our DFT results to be only an approximation of the true spectra of the original input signals prior to digital sampling. Although there are ways to minimize leakage, we can't eliminate it entirely. Thus, we need to understand exactly what effect it has on our DFT results. Let's start from the beginning. DFTs are constrained to operate on a finite set of N input values sampled at a sample rate of f5, to produce an N-point transform whose discrete outputs are associated with the individual analytical frequencies f analysis (m), with (eqn. 24) Equation (eqn. 24), illustrated in DFT Example 1, may not seem like a problem, but it is. The DFT produces correct results only when the input data sequence contains energy precisely at the analysis frequencies given in (eqn. 24), at integral multiples of our fundamental frequency f,/ N. If the input has a signal component at some intermediate frequency between our analytical frequencies of mfs /N, say 1.5fs /N, this input signal will show up to some degree in all of the N Output analysis frequencies of our DFT! (We typically say that input signal energy shows up in all of the DFT's output bins, and we'll see, in a moment, why the phrase "output bins" is appropriate.) Let's understand the significance of this problem with another DFT example.
Assume we're taking a 64-point D.FT of the sequence indicated by the dots in FIG. 7(a). The sequence is a sinewave with exactly three cycles contained in our N = 64 samples. FIG. 7(b) shows the first half of the DFT of the input sequence and indicates that the sequence has an average value of zero (X(0) = 0) and no signal components at any frequency other than the m = 3 frequency. No surprises so far. FIG. 7(a) also shows, for example, the m = 4 sinewave analysis frequency, superimposed over the input sequence, to remind us that the analytical frequencies always have an integral number of cycles over our total sample interval of 64 points. The sum of the products of the input sequence and the m = 4 analysis frequency is zero. (Or we can say, the correlation of the input sequence and the m = 4 analysis frequency is zero.) The sum of the products of this particular three-cycle input sequence and any analysis frequency other than m = 3 is zero. Continuing with our leakage example, the dots in FIG. 8(a) show an input sequence having 3.4 cycles over our N = 64 samples. Because the input sequence does not have an integral number of cycles over our 64-sample interval, input energy has leaked into all the other DFT output bins as shown in FIG. 8(b). The nt = 4 bin, for example, is not zero because the sum of the products of the input sequence and the nt = 4 analysis frequency is no longer zero. This is leakage-it causes any input signal whose frequency is not exactly at a DFT bin center to leak into all of the other DFT output bins. Moreover, leakage is an unavoidable fact of life when we perform the DFT on real-world finite length time sequences. Now, as the English philosopher Douglas Adams would say, "Don't panic." Let's take a quick look at the cause of leakage to learn how to predict and minimize its unpleasant effects. To understand the effects of leakage, we need to know the amplitude response of a DFT when the DFT's input is an arbitrary, real sinusoid. Although Sections 3.14 and 3.15 discuss this issue in de tail, for our purposes, here, we'll just say that, for a real cosine input having k cycles in the N-point input time sequence, the amplitude response of an N-point DFT bin in terms of the bin index nt is approximated by the sine function (eqn. 25) FIG. 9 DFT positive frequency response due to an N-point input sequence containing k cycles of a real cosine: (a) amplitude response as a function of bin index m; (b) magnitude response as a function of frequency in Hz, We'll use (eqn. 25), illustrated in FIG. 9(a), to help us determine how much leakage occurs in DFTs. We can think of the curve in FIG. 9(a), comprising a main lobe and periodic peaks and valleys known as side-lobes, as the continuous positive spectrum of an N-point, real cosine time sequence having k complete cycles in the N-point input time interval. The DFT's out puts are discrete samples that reside on the curves in FIG. 9; that is, our DFT output will be a sampled version of the continuous spectrum. (We show the DFT's magnitude response to a real input in terms of frequency (Hz) in FIG. 9(b).) When the DFT's input sequence has exactly an integral k number of cycles (centered exactly in the m = k bin), no leakage occurs, as in FIG. 9, because when the angle in the numerator of (eqn. 25) is a nonzero integral multiple of it, the sine of that angle is zero. By way of example, we can illustrate again what happens when the input frequency k is not located at a bin center. Assume that a real 8-kHz sinusoid, having unity amplitude, has been sampled at a rate of f,, = 32,000 samples/s. If we take a 32-point DFT of the samples, the DFT's frequency resolution, or bin spacing, is fs /N -= 32,000/32 Hz --, 1.0 kHz. We can predict the DFT's magnitude response by centering the input sinusoid's spectral curve at the positive frequency of 8 kHz, as shown in FIG. 10(a). The dots show the DFT's output bin magnitudes.
Again, here's the important point to remember: the DFT output is a sampled version of the continuous spectral curve in FIG. 10(a). Those sampled values in the frequency-domain, located at mf,/N, are the dots in FIG. 10(a). Because the input signal frequency is exactly at a DFT bin center, the DFT results have only one nonzero value. Stated in another way, when an input sinusoid has an integral number of cycles over N time-domain input sample values, the DFT outputs reside on the continuous spectrum at its peak and exactly at the curve's zero crossing points. From (eqn. 25) we know the peak output magnitude is 32/2 = 16. (If the real input sinusoid had an amplitude of 2, the peak of the response curve would be 2 32/2, or 32.) FIG. 10(b) illustrates DFT leakage where the input frequency is 8.5 kHz, and we see that the frequency-domain sampling results in nonzero magnitudes for all DFT output bins. An 8.75-kHz input sinusoid would result in the leaky DFT output shown in FIG. 10(c). If we're sitting at a computer studying leak age by plotting the magnitude of DFT output values, of course, we'll get the dots in FIG. 10 and won't see the continuous spectral curves. At this point, the attentive reader should be thinking: "If the continuous spectra that we're sampling are symmetrical, why does the DFT output in FIG. 8(b) look so asymmetrical?" In FIG. 8(b), the bins to the right of the third bin are decreasing in amplitude faster than the bins to the left of the third bin. "And another thing, evaluating the continuous spectrum's X(mf,) function at an abscissa value of 0.4 gives a magnitude scale factor of 0.75. Applying this factor to the DFT's maximum possible peak magnitude of 32, we should have a third bin magnitude of approximately 32 • 0.75 = 24-but FIG. 8(b) shows that the third-bin magnitude is slightly greater than 25. What's going on here?" We answer this by remembering what FIG. 8(b) really represents. When examining a DFT out put, we're normally interested only in the m = 0 to ni = (N/2-1) bins. Thus, for our 3.4 cycles per sample interval example in FIG. 8(b), only the first 32 bins are shown. Well, the DFT is periodic in the frequency domain as illustrated in FIG. 11. (We address this periodicity issue in Section 17.) Upon examining the DFT's output for higher and higher frequencies, we end up going in circles, and the spectrum repeats itself forever.
The more conventional way to view a DFT output is to unwrap the spectrum in FIG. 11 to get the spectrum in FIG. 12. FIG. -12 shows some of the additional replications in the spectrum for the 3.4 cycles per sample interval example. Concerning our DFT output asymmetry problem, as some of the input 3.4-cycle signal amplitude leaks into the 2nd bin, the 1st bin, and the 0th bin, leakage continues into the -1st bin, the -2nd bin, the -3rd bin, etc. Remember, the 63rd bin is the -1st bin, the 62nd bin is the -2nd bin, and so on. These bin equivalencies allow us to view the DFT output bins as if they ex tend into the negative frequency range, as shown in FIG. 13(a). The result is that the leakage wraps around the m = 0 frequency bin, as well as around the in = N frequency bin. This is not surprising, because the m = 0 frequency is the m = N frequency. The leakage wraparound at the m = 0 frequency accounts for the asymmetry about the DFT's in = 3 bin in FIG. 8(b).
Recall from the DFT symmetry discussion, when a DFT input sequence x(n) is real, the DFT outputs from in = 0 to in = (N/2-1) are redundant with frequency bin values for in > (N/2), where N is the DFT size. The nith DFT output will have the same magnitude as the (N-ni)th DFT output. That is, I X(m) I = I X(N- m) I I. What this means is that leakage wraparound also occurs about the in N/2 bin. This can be illustrated using an input of 28.6 cycles per sample interval (32 - 3.4) whose spectrum is shown in FIG. 13(b). Notice the similarity between FIG. 13(a) and FIG. 13(6). So the DFT exhibits leakage wraparound about the ni = 0 and ni = N/2 bins. Minimum leakage asymmetry will occur near the N/4th bin as shown in FIG. 14(a) where the full spectrum of a 16.4 cycles per sample interval input is provided. FIG. 14(b) shows a close-up view of the first 32 bins of the 16.4 cycles per sample interval spectrum. You could read about leakage all day. However, the best way to appreciate its effects is to sit down at a computer and use a software program to take DFTs, in the form of fast Fourier transforms (FFTs), of your personally generated test signals like those in Figures 3-7 and 3-8. You can, then, experiment with different combinations of input frequencies and various DFT sizes. You'll be able to demonstrate that DFT leakage effect is troublesome because the bins containing low-level signals are corrupted by the side-lobe levels from neighboring bins containing high-amplitude signals. Although there's no way to eliminate leakage completely, an important technique known as windowing is the most common remedy to reduce its unpleasant effects. Let's look at a few DFT window examples.
9. Windowing reduces DFT leakage by minimizing the magnitude of (eqn. 25)'s sinc function's sin(x)/x sidelobes shown in FIG. 9. We do this by forcing the amplitude of the input time sequence at both the beginning and the end of the sample interval to go smoothly toward a single common amplitude value. FIG. 15 shows how this process works. If we consider the infinite-duration time signal shown in FIG. 15(a), a DFT can only be per formed over a finite-time sample interval like that shown in FIG. 15(c). We can think of the DFT input signal in FIG. 15(c) as the product of an input signal existing for all time, FIG. 15(a), and the rectangular window whose magnitude is 1 over the sample interval shown in FIG. 15(b). Any time we take the DFT of a finite-extent input sequence we are, by default, multiplying that sequence by a window of all ones and effectively multiplying the input values outside that window by zeros. As it turns out, (eqn. 25)'s sinc function's sin(x)/x shape, shown in FIG. 9, is caused by this rectangular window because the continuous Fourier transform of the rectangular window in FIG. 15(b) is the sinc function. As we'll soon see, it's the rectangular window's abrupt changes be tween one and zero that are the cause of the sidelobes in the sin(x)/x sinc function. To minimize the spectral leakage caused by those sidelobes, we have to reduce the sidelobe amplitudes by using window functions other than the rectangular window. Imagine if we multiplied our DFT input, FIG. 15(c), by the triangular window function shown in FIG. 15(d) to obtain the windowed input signal shown in FIG. 15(e). Notice that the values of our final input signal appear to be the same at the beginning and end of the sample interval in FIG. 15(e). The reduced discontinuity decreases the level of relatively high frequency components in our overall DFT output; that is, our DFT bin sidelobe levels are reduced in magnitude using a triangular window. There are other window functions that reduce leakage even more than the triangular window, such as the Hanning window in FIG. 15(f). The product of the window in FIG. 15(f) and the input sequence provides the signal shown in FIG. 15(g) as the input to the DFT. Another common window function is the Hamming window shown in FIG. 15(h). It's much like the Hanning window, but it's raised on a pedestal. Before we see exactly how well these windows minimize DFT leakage, let's define them mathematically. Assuming that our original N input signal samples are indexed by n, where 0 n N-1, we'll call the N time-domain window coefficients w(n); that is, an input sequence x(n) is multiplied by the corresponding window w(n) coefficients before the DFT is performed. So the DFT of the windowed x(n) input sequence, X,v (m), takes the form of (eqn. 26) To use window functions, we need mathematical expression of them in terms of n. The following expressions define our window function coefficients: Rectangular window: (also called the uniform or boxcar, win (low) Triangular window: (very similar to the Bartlett, and l'arzen windows) Harming window: (also called the raised cosine, Hann, or von f Win window) Hamming window: (eqn. 27) (eqn. 28) (eqn. 29) (eqn. 30) If we plot the w(n) values from Eqs. (eqn. 27) through (eqn. 30), we'd get the corresponding window functions like those in Figures 15(b), 15(d), 3-15( 1 ), and 15(h). [In the literature, the equations for window functions depend on the range of the sample index n. We define n to be in the range 0 < n < N-1. Some authors define n to be in the range -N/2 < n < N/2, in which case, for example, the expression for the I 'arming window would have a sign change and be w(n)= 0.5 + 0.5cos(2 pi n/N-1). ]
The rectangular window's amplitude response is the yardstick we normally use to evaluate another window function's amplitude response; that is, we typically get an appreciation for a window's response by comparing it to the rectangular window that exhibits the magnitude response shown in FIG. 9(b). The rectangular window's sin(x)/x magnitude response, I W(nz) I, is repeated in FIG. 16(a). Also included in FIG. 16(a) are the Hamming, Hanning, and triangular window magnitude responses. (The frequency axis in FIG. 16 is such that the curves show the response of a single N-point DFT bin when the various window functions are used.) We can see that the last three windows give reduced sidelobe levels relative to the rectangular window. Because the Hamming, Harming, and triangular windows reduce the time-domain signal levels applied to the DFT, their main lobe peak values are reduced relative to the rectangular window. (Because of the near-zero w(n) coefficients at the beginning and end of the sample interval, this signal level loss is called the processing gain, or loss, of a window.) Be that as it may, we're primarily interested in the windows' sidelobe levels, which are difficult to see in FIG. 16(a)'s linear scale. We will avoid this difficulty by plotting the windows' magnitude responses on a logarithmic decibel scale, and normalize each plot so its main lobe peak values are zero dB. ( Section E provides a discussion of the origin and utility of measuring frequency-domain responses on a logarithmic scale using decibels.) Defining the log magnitude response to be I Wm(m) I, we get I W(R(m) I by using the expression (eqn. 31) (The I W(0)| term in the denominator of (eqn. 31) is the value of W(m) at the peak of the main lobe when m = 0.) The I WdB(m) I curves for the various window functions are shown in FIG. 16(b). Now we can really see how the various window sidelobe responses compare to each other. Looking at the rectangular window's magnitude response we see that its main lobe is the most narrow, fs /N. However, unfortunately, its first sidelobe level is only -13 dB below the Main lobe peak, which is not so good. (Notice that we're only showing the positive frequency portion of the window responses in FIG. 16.) The triangular window has reduced sidelobe levels, but the price we've paid is that the triangular window's main lobe width is twice as wide as that of the rectangular window's. The various nonrectangular windows' wide main lobes degrade the windowed DFT's frequency resolution by almost a factor of two. However, as we'll see, the important benefits of leakage reduction usually outweigh the loss in DFT frequency resolution. Notice the further reduction of the first sidelobe level, and the rapid sidelobe roll-off of the Harming window. The Hamming window has even lower first sidelobe levels, but this window's sidelobes roll off slowly relative to the Hanning window. This means that leakage three or four bins away. From the center bin is lower for the Hamming window than for the Nanning window, and leakage a half dozen or so bins away from the center bin is newer for the Hanning window than for the Hamming window. When we apply the Hanning window to FIG. 8(a)'s 3.4 cycles per sample interval example, we end up with the DFT input shown in Figure POi -1 7(a) under the Hanning window envelope. The DIT outputs for the windowed waveform are shown in FIG. 17(b) along with the DFT results with no windowing, i.e., the rectangular window. As we expected, the shape of the Hanning window's response looks broader and has a lower peak amplitude, but its sidelobe leakage is noticeably reduced from that of the rectangular window.
We can demonstrate the benefit of using a window function to help us detect a low-level signal in the presence of a nearby high-level signal. Let's add 64 samples of a 7 cycles per sample interval sinewave, with a peak amplitude of only 0.1, to FIG. 8(a)'s unity-amplitude 3.4 cycles per sample sinewave. When we apply a Harming window to the sum of these sinewaves, we get the time-domain input shown in FIG. 18(a). Had we not windowed the input data, our DFT output would be the squares in FIG. 18(b) where DFT leakage causes the input signal component at m = 7 to be barely discernible. However, the DFT of the windowed data shown as the triangles in FIG. 18(b) makes it easier for us to detect the presence of the m = 7 signal component. From a practical standpoint, people who use the DFT to per form real-world signal detection have learned that their overall frequency resolution and signal sensitivity are affected much more by the size and shape of their window function than the mere size of their DFTs. As we become more experienced using window functions on our DFT input data, we'll see how different window functions have their own individual advantages and disadvantages. Furthermore, regardless of the window function used, we've decreased the leakage in our DFT output from that of the rectangular window. There are many different window functions de- scribed in the literature of digital signal processing--so many, in fact, that they've been named after just about everyone in the digital signal processing business. It's not that clear that there's a great deal of difference among many of these window functions. What we find is that window selection is a trade off between main lobe widening, first sidelobe levels, and how fast the side lobes decrease with increased frequency. The use of any particular window depends on the application [5], and there are many applications. Windows are used to improve DFT spectrum analysis accuracy[6], to de sign digital filters [7,8], to simulate antenna radiation patterns, and even in the hardware world to improve the performance of certain mechanical force to voltage conversion devices. So there's plenty of window information avail able for those readers seeking further knowledge. (The mother of all technical papers on windows is that by Harris. A useful paper by Nuttall corrected and extended some portions of Harris's paper.) Again, the best way to appreciate windowing effects is to have access to a computer software package that contains DFT, or FFT, routines and start analyzing windowed signals. (By the way, while we delayed their discussion until Section 5.3, there are two other commonly used window functions that can be used to reduce DFT leak age. They're the Chebyshev and Kaiser window functions, which have adjustable parameters, enabling us to strike a compromise between widening main lobe width and reducing sidelobe levels.) [Perhaps FIG. 19(a) is why individual DFT outputs are called "bins." Any signal energy under a sin(x)/x curve will show up in the enclosed storage compartment of that DFT's output sample. ]
10. Scalloping is the name used to describe fluctuations in the overall magnitude response of an N-point DFT. Although we derive this fact in Section 16, for now we'll just say that when no input windowing function is used, the sin(x)/x shape of the sinc function's magnitude response applies to each DFT output bin. FIG. 19(a) shows a DFT's aggregate magnitude response by superimposing several sin(x)/x bin magnitude responses. (Because the sinc function's sidelobes are not key to this discussion, we don't show them in FIG. 19(a).) Notice from FIG. 19(b) that the overall DFT frequency domain response is indicated by the bold envelope curve. This rippled curve, also called the picket fence effect, illustrates the processing loss for input frequencies between the bin centers. From FIG. 19(b), we can determine that the magnitude of the DFT response fluctuates from 1.0, at bin center, to 0.637 halfway between bin centers. If we're interested in DFT output power levels, this envelope ripple exhibits a scalloping loss of almost -4 dB halfway between bin centers. FIG. 19 illustrates a DFT output when no window (i.e., a rectangular window) is used. Because nonrectangular window functions broaden the DFT's main lobe, their use results in a scalloping loss that will not be as severe as the rectangular window. That is, their wider main lobes overlap more and fill in the valleys of the envelope curve in FIG. 19(b). For example, the scalloping loss of a Hanning window is approximately 0.82, or -1.45 dB, halfway between bin centers. Scalloping loss is not, however, a severe problem in practice. Real-world signals normally have bandwidths that span many frequency bins so that DFT magnitude response ripples can go almost unnoticed. Let's look at a scheme called zero padding that's used to both alleviate scalloping loss effects and to improve the DFT's frequency resolution. 11. One popular method used to improve DFT spectral estimation is known as zero padding. This process involves the addition of zero-valued data samples to an original DFT input sequence to increase the total number of input data samples. Investigating this zero padding technique illustrates the DFT's important property of frequency-domain sampling alluded to in the discussion on leakage. When we sample a continuous time-domain function, having a continuous Fourier transform (CFT), and take the DFT of those samples, the DFT results in a frequency-domain sampled approximation of the CFT. The more points in our DFT, the better our DFT output approximates the CFT. To illustrate this idea, suppose we want to approximate the CFT of the continuous f(t) function in FIG. 20(a). This f(t) waveform extends to infinity in both directions but is nonzero only over the time interval of T seconds. If the nonzero portion of the time function is a sinewave of three cycles in T seconds, the magnitude of its CFT is shown in FIG. 20(b). (Because the CFT is taken over an infinitely wide time interval, the CFT has infinitesimally small frequency resolution, resolution so fine-grained that it's continuous.) It's this CFT that we'll approximate with a DFT.
Suppose we want to use a 16-point DFT to approximate the CFT of f(t) in FIG. 20(a). The 16 discrete samples of f(t), spanning the three periods of f(t)'s sinusoid, are those shown on the left side of FIG. 21(a). Applying those time samples to a 16-point DFT results in discrete frequency-domain samples, the positive frequency of which are represented by the dots on the right side of FIG. 21(a). We can see that the DFT output samples FIG. 20(b)'s CFT. If we append (or zero pad) 16 zeros to the input sequence and take a 32-point DFT, we get the output shown on the right side of FIG. 21(b), where we've increased our DFT frequency sampling by a factor of two. Our DFT is sampling the input function's CFT more often now. Adding 32 more zeros and taking a 64-point DFT, we get the output shown on the right side of FIG. 21 (c). The 64-point DFT output now begins to show the true shape of the CFT. Adding 64 more zeros and taking a 128-point DFT, we get the output shown on the right side of FIG. 21(d). The DFT frequency domain sampling characteristic is obvious now, but notice that the bin index for the center of the main lobe is different for each of the DFT outputs in FIG. 21. Does this mean we have to redefine the DFT's frequency axis when using the zero-padding technique? Not really. If we perform zero padding on L nonzero input samples to get a total of N time samples for an N-point DFT, the zero-padded DFT output bin center frequencies are related to the original L by our old friend (eqn. 5), or center frequency of the m (eqn. 32) So in our FIG. 21(a) example, we use (eqn. 32) to show that, although the zero-padded DFT output bin index of the main lobe changes as N in creases, the zero-padded DFT output frequency associated with the main lobe remains the same. The following list shows how this works: Do we gain anything by appending more zeros to the input sequence and taking larger DFTs? Not really, because our 128-point DFT is sampling the input's CFT sufficiently now in FIG. 21(d). Sampling it more often with a larger DFT won't improve our understanding of the input's frequency content. The issue here is that adding zeros to an input sequence will improve our DFT's output resolution, but there's a practical limit on how much we gain by adding more zeros. For our example here, a 128-point DFT shows us the detailed content of the input spectrum. We've hit a law of diminishing re turns here. Performing a 256-point or 512-point DFT, in our case, would serve little purpose. There's no reason to oversample this particular input sequence's CFT. Of course, there's nothing sacred about stopping at a 128-point DFT. Depending on the number of samples in some arbitrary input sequence and the sample rate, we might, in practice, need to append any number of zeros to get some desired DFT frequency resolution. Notice that the DFT sizes (N) we've discussed are powers of 2 (64, 128, 256, 512). That's be cause we actually perform DFTs using a special algorithm known as the fast Fourier transform (FFT). As we'll see in Section 4, the typical implementation of the FFT requires that N be a power of 2. There are two final points to be made concerning zero padding. First, the DFT magnitude expressions in Eqs. (eqn. 17) and (eqn. 17') don't apply if zero padding is being used. If we perform zero padding on L nonzero samples of a sinusoid whose frequency is located at a bin center to get a total of N input samples for an N-point DFT, we must replace the N with L in Eqs. (eqn. 17) and (eqn. 17') to predict the DFT's output magnitude for that particular sinewave. Second, in practical situations, if we want to perform both zero padding and windowing on a sequence of input data samples, we must be careful not to apply the window to the entire input including the appended zero-valued samples. The window function must be applied only to the original nonzero time samples, otherwise the padded zeros will zero out and distort part of the window function, leading to erroneous results. (Section 4.5 gives additional practical pointers on performing the DFT using the HI algorithm to analyze real-world signals.) To digress slightly, now's a good time to define the term discrete-time Fourier transform (DTFT) that the reader may encounter in the literature. The DTFT is the continuous Fourier transform of an L-point discrete time domain sequence; and some authors use the DTFT to describe many of the digital signal processing concepts we've covered in this section. On a computer we can't perform the DTFT because it has an infinitely fine frequency resolution-but we can approximate the DTFT by performing an N-point DFT on an L-point discrete time sequence where N > L. That is, in fact, what we did in FIG. 21 when we zero-padded the original 16-point time sequence. (When N = L the DTFT approximation is identical to the DFT.) To make the connection between the DTFT and the DFT, know that the infinite-resolution DTFT magnitude (i.e., continuous Fourier transform magnitude) of the 16 non-zero time samples in FIG. 21(a) is the shaded sin(x)/x-like spectral function in FIG. 21. Our DFTs approximate (sample) that function. Increased zero padding of the 16 non-zero time samples merely interpolates our DFT's sampled version of the DTFT function with smaller and smaller frequency-domain sample spacing. Please keep in mind, however, that zero padding does not improve our ability to resolve, to distinguish between, two closely spaced signals in the frequency domain. (For example, the main lobes of the various spectra in FIG. 21 do not change in width, if measured in Hz, with increased zero padding.) To improve our true spectral resolution of two signals, we need more non-zero time samples. The rule by which we must live is: to realize Hz spectral resolution, we must collect 1/F seconds worth of non-zero time samples for our DFT processing. We'll discuss applications of time-domain zero padding in Section 15, revisit the DTFT in Section 17, and frequency-domain zero padding in Section 28. 12. There are two types of processing gain associated with DFTs. People who use the DFT to detect signal energy embedded in noise often speak of the DFT's processing gain because the DFT can pull signals out of background noise. This is due to the inherent correlation gain that takes place in any N-point DFT. Be yond this natural processing gain, additional integration gain is possible when multiple DFT outputs are averaged. Let's look at the DFT's inherent processing gain first.
12.1 The concept of the DFT having processing gain is straightforward if we think of a particular DFT bin output as the output of a narrowband filter. Because a DFT output bin has the amplitude response of the sin(x)/x function, that bin's output is primarily due to input energy residing under, or very near, the bin's main lobe. It's valid to think of a DFT bin as a kind of bandpass filter whose band center is located at mL/N. We know from (eqn. 17) that the maximum possible DFT output magnitude increases as the number of points (N) in a DFT increases. Also, as N increases, the DFT output bin main lobes become more narrow. So a DFT output bin can be treated as a bandpass filter whose gain can be increased and whose bandwidth can be reduced by increasing the value of N. Decreasing a bandpass filter's bandwidth is useful in energy detection because the frequency resolution improves in addition to the filter's ability to minimize the amount of background noise that resides within its passband. We can demonstrate this by looking at the DFT of a spectral tone (a constant frequency sinewave) added to random noise. FIG. 22(a) is a logarithmic plot showing the first 32 outputs of a 64-point DFT when the input tone is at the center of the DFT's m = 20th bin. The output power levels (DFT magnitude squared) in FIG. 22(a) are normalized so that the highest bin output power is set to 0 dB. Because the tone's original signal power is below the average noise power level, the tone is a bit difficult to detect when N = 64. (The time-domain noise, used to generate FIG. 22(a), has an average value of zero, i.e., no DC bias or amplitude off set.) If we quadruple the number of input samples and increase the DFT size to N = 256, we can now see the tone power raised above the average background noise power as shown for nt = 80 in FIG. 22(b). Increasing the DFT's size to N = 1024 provides additional processing gain to pull the tone further up out of the noise as shown in FIG. 22(c).
To quantify the idea of DFT processing gain, we can define a signal-to noise ratio (SNR) as the DFT's output signal-power level over the average output noise-power level. (In practice, of course, we like to have this ratio as large as possible.) For several reasons, it's hard to say what any given single DFT output SNR will be. That's because we can't exactly predict the energy in any given N samples of random noise. Also, if the input signal frequency is not at bin center, leakage will raise the effective background noise and reduce the DFT's output SNR. In addition, any window being used will have some effect on the leakage and, thus, on the output SNR. What we'll see is that the DFT's output SNR increases as N gets larger because a DFT bin's output noise standard deviation (rms) value is proportional to 4f , and the DFT's output magnitude for the bin containing the signal tone is proportional to IV. More generally for real inputs, if N> N', an N-point DFT's output SNR N increases over the N'-point DFT SNR N, by the following relationship: (eqn. 33) If we increase a DFT's size from N' to N = 2N', from (eqn. 33), the DFT's out put SNR increases by 3 dB. So we say that a DFT's processing gain increases by 3 dB whenever N is doubled. Be aware that we may double a DFT's size and get a resultant processing gain of less than 3 dB in the presence of random noise; then again, we may gain slightly more than 3 dB. That's the nature of random noise. If we perform many DFTs, we'll see an average processing gain, shown in FIG. 23(a), for various input signal SNRs. Because we're interested in the slope of the curves in FIG. 23(a), we plot those curves on a logarithmic scale for N in FIG. 23 (3) where the curves straighten out and become linear. Looking at the slope of the curves in FIG. 23(b), we can now see the 3 dB increase in processing gain as N doubles so long as N is greater than 20 or 30 and the signal is not overwhelmed by noise. There's nothing sacred about the absolute values of the curves in Figures 23(a) and 23(b). They were generated through a simulation of noise and a tone whose frequency was at a DFT bin center. Had the tone's frequency been between bin centers, the processing gain curves would have been shifted downward, but their shapes would still be the same, i that is, (eqn. 33) is still valid regardless of the input tone's frequency. 12.2 Theoretically, we could get very large DFT processing gains by increasing the DFT size arbitrarily. The problem is that the number of necessary DFT multiplications increases proportionally to N 2 , and larger DFTs become very computationally intensive. Because addition is easier and faster to perform than multiplication, we can average the outputs of multiple DFTs to obtain further processing gain and signal detection sensitivity. The subject of averaging multiple DFT outputs is covered in Section 11.3. The curves would be shifted downward, indicating a lower SNI, because leakage would raise the average noise-power level, and scalloping loss would reduce the l)11 . bin's output power level_ 13. We conclude this section by providing the mathematical details of two important aspects of the DFT. First, we obtain the expressions for the DFT of a rectangular function (rectangular window), and then we'll use these results to illustrate the magnitude response of the DFT. We're interested in the DFT's magnitude response because it provides an alternate viewpoint to under stand the leakage that occurs when we use the DFT as a signal analysis tool. One of the most prevalent and important computations encountered in digital signal processing is the DFT of a rectangular function. We see it in sampling theory, window functions, discussions of convolution, spectral analysis, and in the design of digital filters. As common as it is, however, the literature covering the DFT of rectangular functions can be confusing for several reasons for the digital signal processing beginner. The standard mathematical notation is a bit hard to follow at first, and sometimes the equations are presented with too little explanation. Compounding the problem, for the beginner, are the various expressions of this particular DFT. In the literature, we're likely to find any one of the following forms for the DFT of a rectangular function: (eqn. 34) In this section we'll show how the forms in (eqn. 34) were obtained, see how they're related, and create a kind of Rosetta stone table allowing us to move back and forth between the various DFT expressions. Take a deep breath and let's begin our discussion with the definition of a rectangular function.
13.1 A general rectangular function x(n) can be defined as N samples containing K unity-valued samples as shown in FIG. 24. The full N-point sequence, x(n), is the rectangular function that we want to transform. We call this the general form of a rectangular function because the K unity samples begin at a arbitrary index value of -n0 . Let's take the DFT of x(n) in FIG. 24 to get our desired X(m). Using m as our frequency-domain sample index, the expression for an N-point DFT is (eqn. 35) With x(n) being nonzero only over the range of -n. n -no + (K-1), we can modify the summation limits of (eqn. 35) to express X(m) as (eqn. 36) because only the K samples contribute to X(m). That last step is important be cause it allows us to eliminate the x(n) terms and make (eqn. 36) easier to handle. To keep the following equations from being too messy, let's use the dummy variable q = 27cm / N. OK, here's where the algebra comes in. Over our new limits of summation, we eliminate the factor of 1 and (eqn. 36) becomes (eqn. 37) The series inside the brackets of (eqn. 37) allows the use of a summation, such as (eqn. 38) Equation (eqn. 38) certainly doesn't look any simpler than (eqn. 36), but it is. Equation (eqn. 38) is a geometric series and, from the discussion in Section B, it can be evaluated to the closed form of (eqn. 39) We can now simplify (eqn. 39)---here's the clever step. If we multiply and divide the numerator and denominator of (eqn. 39)'s right-hand side by the appropriate half-angled exponentials, we break the exponentials into two parts and get (eqn. 40) Let's pause for a moment here to remind ourselves where we're going. We're trying to get (eqn. 40) into a usable form because it's part of (eqn. 38) that we're using to evaluate X(m) in (eqn. 36) in our quest for an understandable expression for the DFT of a rectangular function. Equation (eqn. 40) looks even more complicated than (eqn. 39), but things can be simplified inside the parentheses. From Euler's equation: sin(e) = (ei° - e-1 )/2j, (eqn. 40) becomes (eqn. 41) Substituting (eqn. 41) for the summation in (eqn. 38), our expression for X(q) becomes (eqn. 42) Returning our dummy variable q to its original value of 2 pi m/N, (eqn. 43) So there it is (Whew!). Equation (eqn. 43) is the general expression for the DFT of the rectangular function as shown in FIG. 24. Our X(m) is a complex expression (pun intended) where a ratio of sine terms is the amplitude of X(m) and the exponential term is the phase angle of X(m). The ratio of sines factor in (eqn. 43) lies on the periodic curve shown in FIG. 25(a), and like all N-point DFT representations, the periodicity of X(m) is N. This curve is known as the Dirichlet kernel (or the aliased sine function) and has been thoroughly de scribed in the literature. (It's named after the nineteenth-century German mathematician Peter Dirichlet (pronounced dee-ree-klay), who studied the convergence of trigonometric series used to represent arbitrary functions.) We can zoom in on the curve at the ni = 0 point and see more detail in FIG. 25(b). The dots are shown in FIG. 25(b) to remind us that the DFT of our rectangular function results in discrete amplitude values that lie on the curve. So when we perform DFTs, our discrete results are sampled values of the continuous sine function's curve in FIG. 25(a). As we'll show later, we're primarily interested in the absolute value, or magnitude, of the Dirichlet kernel in (eqn. 43). That magnitude I X(m)I is shown in FIG. 25(c). Although we first saw the sinc function's curve in FIG. 9 in Section 8, where we introduced the topic of DET leakage, we'll encounter this curve often in our study of digital signal processing. [N was an even number in FIG. 24 depicting the x(n). Had N been an odd number, the limits on the summation in (eqn. 35) would have been -(N-1 )/2 n (N-1)/2. Using these alternate limits would have led us to exactly the same X(m) as in (eqn. 43). L’Hopital is pronounced 'lo-pe-tol’, like baby doll. ] For now, there are just a few things we need to keep in mind concerning the Dirichlet kernel. First, the DFT of a rectangular function has a main lobe, centered about the in = 0 point. The peak amplitude of the main lobe is K. This peak value makes sense, right? The ni = 0 sample of a DFT X(0) is the sum of the original samples, and the sum of K unity-valued samples is K. We can show this in a more substantial way by evaluating (eqn. 43) for m O. A difficulty arises when we plug ni = 0 into (eqn. 43) because we end up with sin(0)/sin(0), which is the indeterminate ratio 0/0. Well, hardcore mathematics to the rescue here. We can use L'Hopital's rule to take the derivative of the numerator and the denominator of (eqn. 43), and then set m = 0 to determine the peak value of the magnitude of the Dirichlet kernel." We proceed as (eqn. . 44) which is what we set out to show. (We could have been clever and evaluated (eqn. 35) with m = 0 to get the result of (eqn. 44). Try it, and keep in mind that el° = 1.) Had the amplitudes of the nonzero samples of x(n) been different than unity, say some amplitude A., then, of course, the peak value of the Dirichlet kernel would be A.K instead of just K. The next important thing to notice about the Dirichlet kernel is the main lobe's width. The first zero crossing of (eqn. 43) occurs when the numerator's argument is equal to n. That is, when nmK/N = n. So the value of m at the first zero crossing is given by (eqn. 45) as shown in FIG. 25(b). Thus the main lobe width 2N/K, as shown in FIG. 25(c), is inversely proportional to K. Notice that the main lobe in FIG. 25(a) is surrounded by a series of oscillations, called sidelobes, as in FIG. 25(c). These sidelobe magnitudes decrease the farther they're separated from the main lobe. However, no matter how far we look away from the main lobe, these sidelobes never reach zero magnitude--and they cause a great deal of heartache for practitioners in digital signal processing. These sidelobes cause high-amplitude signals to overwhelm and hide neighboring low-amplitude signals in spectral analysis, and they complicate the design of digital filters. As we'll see in Section 5, the unwanted ripple in the passband and the poor stopband attenuation in simple digital filters are caused by the rectangular function's DFT sidelobes. (The development, analysis, and application of window functions came about to minimize the ill effects of those sidelobes in FIG. 25.) Let's demonstrate the relationship in (eqn. 45) by way of a simple but concrete example. Assume that we're taking a 64-point DFT of the 64-sample rectangular function, with eleven unity values, shown in FIG. 26(a). In this example, N = 64 and K = 11. Taking the 64-point DFT of the sequence in FIG. 26(a) results in an X(m) whose real and imaginary parts, Xrc, a1 (m) and Xima ,(m), are shown in FIG. 26(b) and FIG. 26(c) respectively. FIG. S(b) is a good illustration of how the real part of the DFT of a real input sequence has even symmetry, and FIG. 26(c) confirms that the imaginary part of the DFT of a real input sequence has odd symmetry. Although Xwai (m) and Xini ,g(m) tell us everything there is to know about the DFT of x(n), it's a bit easier to comprehend the true spectral nature of X(ni) by viewing its absolute magnitude. This magnitude, from (eqn. 7), is provided in FIG. 27(a) where the main and sidelobes are clearly evident now. As we expected, the peak value of the main lobe is 11 because we had K = ri samples in x(n). The width of the main lobe from (eqn. 45) is 64/11, or 5.82. Thus, the first positive frequency zero-crossing location lies just below the ni = 6 sample of our discrete 1 X(m) I represented by the squares in FIG. 27(a). The phase angles associated with 1X(m)1, first introduced in Equations (eqn. 6) and (eqn. 8), are shown in FIG. 27(b). To understand the nature of the DFT of rectangular functions more fully, let's discuss a few more examples using less general rectangular functions that are more common in digital signal processing than the x(n) in FIG. 24. FIG. 26 DFT of a rectangular function: (a) original function x(n) ; (b) real part of the DFT of x(n), Xregi (m): (c) imaginary part of the DFT of x(n), Ximag (m). FIG. 27 DFT of a generalized rectangular function: (a) magnitude I x(m)1 ; (b) phase angle in radians. 13.2 DFT of a Symmetrical Rectangular Function Equation (eqn. 43) is a bit complicated because our original function x(n) was so general. In practice, special cases of rectangular functions lead to simpler versions of (eqn. 43). Consider the symmetrical x(n) rectangular function in FIG. 28. As shown in FIG. 28, we often need to determine the DFT of a rectangular function that's centered about the n = 0 index point. In this case, the K unity-valued samples begin at n = -(K-1)/2. So substituting (K-1)/2 for no in (eqn. 43) yields … Because c10 1, (eqn. 46) becomes (eqn. 46) Symmetrical form of the Dirichlet kernel: (eqn. 47) Equation (eqn. 47) indicates that the DFT of the symmetrical rectangular function in FIG. 28 is itself a real function; that is, there's no complex exponential in (eqn. 47), so this particular DFT contains no imaginary part or phase term. As we stated in Section 2, if x(n) is real and even, x(n) = then X ival (nt) is nonzero and XiMag(m) is always zero. We demonstrate this by taking the 64-point DFT of the sequence in FIG. 29(a). Our x(n) is 11 unity-valued samples centered about the n 0 index. Here the DFT results in an X(nt) whose real and imaginary parts are shown in FIG. 29(b) and FIG. 29(c), respectively. As (eqn. 47) predicted, X r,sai (nt) is nonzero and Ximau(m) is zero. The magnitude and phase of X(m) are depicted in FIG. 29(d) and FIG. -29(e). FIG. 28 Rectangular x(n) with K samples centered about n = O. Notice that the magnitudes in FIG. 27(a) and FIG. 29(d) are identical. This verifies the very important shifting theorem of the DFT; that is, the magnitude I X(m) I depends only on the number of nonzero samples in x(n), K, and not on their position relative to the n = 0 index value. Shifting the K unity-valued samples to center them about the n = 0 index merely affects the phase angle of X(m), not its magnitude. FIG. 29 DFT of a rectangular function centered about n = 0: (a) original x(n) (b) Xreal (m); (c) Xmag(m); (d) magnitude of X(m); (e) phase angle of X(m) in radians, FIG. 30 DFT of a symmetrical rectangular function with 31 unity values: (a) original x(n); (b) magnitude of X(m). Speaking of phase angles, it's interesting to realize here that even though Ximag(m) is zero in FIG. 29(c), the phase angle of X(m) is not al ways zero. In this case, X(m)'s individual phase angles in FIG. 29(e) are either +it, zero, or -it radians. With +it and -71 radians both being equal to -1, Xreal (m) X(m) X0(m) … we must. Xreiii (m) is equal to I X(m) I with the signs of I X(m) I 's alternate side lobes reversed.' To gain some further appreciation of how the DFT of a rectangular function is a sampled version of the Dirichlet kernel, let's increase the number of our nonzero x(n) samples. FIG. 30(a) shows a 64-point x(n) … where 31 unity-valued samples are centered about the n = 0 index location. The magnitude of X(ni) is provided in FIG. 30(b). By broadening the x(n) function, i.e., increasing K, we've narrowed the Dirichlet kernel of X(m). This follows from (eqn. 45), right? The kernel's first zero crossing is inversely proportional k) K, so, as we extend the width of K, we squeeze I X(m) I in to ward ni = O. in this example, N 64 and K = 31. From (eqn. 45) the first positive zero crossing of X(m) occurs at 64/31, or just slightly to the right of the m = 2 sample in 'FIG. 30(b). Also notice that the peak value of I X(m) I = K = 31, as mandated by (eqn. 44). t The particular pattern of +n and -71 values in FIG. -29(e) is an artifact of the software used to generate that figure. A different software package may show a different pattern, but as long as the nonzero phase samples are either +ix or -N, the phase results will be correct. Figure 3 31 Rectangular function with N unity-valued samples. 13.3 The DFT of a special form of x(n) is routinely called for, leading to yet another simplified form of (eqn. 43). In the literature, we often encounter a rectangular function where K = N; that is, all N samples of x(n) are nonzero, as shown in FIG. 31. In this case, the N unity-valued samples begin at n = --no = -(N-1)/2. We obtain the expression for the DFT of the function in FIG. 31 by substituting K - N and n 0 = (N-1)/2 in (eqn. 43) to get (eqn. 48) Equation (eqn. 48) takes the first form of (eqn. 34) that we alluded to at the beginning of Section 13.t FIG. 32 demonstrates the meaning of (eqn. 48). The DFT magnitude of the all ones function, x(n) in FIG. 32(a), is shown in FIG. 32(b) and FIG. 32(c). Take note that if m was continuous (eqn. 48) describes the shaded curves in FIG. 32(b) and FIG. 32(c). If m is restricted to be integers, then (eqn. 48) represents the dots in those figures. The Dirichlet kernel of X(m) in FIG. 32(h) is now as narrow as it can get. The main lobe's first positive zero crossing occurs at the m = 64/64 = 1 sample in FIG. 32(b) and the peak value of I X(m) I = N = 64. With x(n) being all ones, I X(m) I is zero for all m *0. The sinc function in (eqn. 48) is of utmost importance-as we'll see at the end of this section, it defines the over all DFT frequency response to an input sinusoidal sequence, and it's also the amplitude response of a single DFT bin. [By the way, there's nothing official about calling (eqn. 48) a Type 1 Dirichlet kernel. We're using the phrase Type 1 merely to distinguish (eqn. 48) from other mathematical expressions for ] FIG. 32 All ones function: (a) rectangular function with N = 64 unity-valued samples: (b) DFT magnitude of the all ones time function: (c) close up view of the DFT magnitude of an all ones time function. FIG. 33 Relationships between an angle a. line a = sin(a), and us chord b: (a) large angle a: (b) small angle a. The form of (eqn. 48) allows us to go one step further to identify the most common expression for the DFT of an all ones rectangular function found in the literature. To do this, we have to use an approximation principle found in the mathematics of trigonometry that you may have heard before. It states that when a is small, then sin(a) is approximately equal to a, i.e., sin(a) a. This idea comes about when we consider a pie-shaped section of a circle whose radius is 1 as shown in FIG. 33(a). That section is defined by the length of the arc a measured in radians and a's chord b. If we draw a right triangle inside the section, we can say that a = sin(a). As a gets smaller the long sides of our triangle become almost parallel, the length of chord b approaches the length of arc a, and the length of line a approaches the length of b. So, as depicted in FIG. 33(b), when a is small, a b a = sin(a). We use this sin(a) a approximation when we look at the denominator of (eqn. 48). When nm/N is small, then sin(nm/N) is approximately equal to nm/N. So we can, when N is large, state All Ones form of the Dirichlet kernel (Type 2): -) sin(rtm) (eqn. 49) It has been shown that when N is larger than, say, 10 in (eqn. 48), (eqn. 49) accurately describes the DFT's output. t Equation (eqn. 49) is often normalized by dividing it by N, so we can express the normalized DFT of an all ones rectangular function as An Ones form of the Dirichlet kernel (Type 3): --o X(m) sin(Em) itm (eqn. 50) Equation (eqn. 50), taking the second form of (eqn. 34) that is so often seen in the literature, also has the DFT magnitude shown in FIG. 32(b) and FIG. 32(c). -I- We can be comfortable with this result because, if we let K = N, we'll see that the peak value of X(m) in (eqn. 49), for m 0, is equal to TV, which agrees with (eqn. 44). 13.4 Time and Frequency Axes Associated with Rectangular Functions Let's establish the physical dimensions associated with the n and m index values. So far in our discussion, the n index was merely an integer enabling us to keep track of individual x(n) sample values. If the n index represents instants in time, we can identify the time period separating adjacent x(n) samples to establish the time scale for the x(n) axis and the frequency scale for the X(m) axis. Consider the time-domain rectangular function given in FIG. 34(a). That function comprises N time samples obtained t, seconds apart, and the full sample interval is Nt, seconds. Each x(n) sample occurs at nt, seconds for some value of n. For example, the n = 9 sample value, x(9) = 0, occurs at 91-, seconds. The frequency axis of X(m) can be represented in a number of different ways. Three popular types of DFT frequency axis labeling are shown in FIG. 34(b) and listed in Table 1. Let's consider each representation individually. FIG. 34 DFT time and frequency axis dimensions: (a) time-domain axis uses time index n; (b) various representations of the DFT's frequency axis. Table 3 1 Characteristics of Various DFT Frequency Axis Representations 13.4.1 DFT Frequency Axis in Hz. If we decide to relate X(m) to the time sample period t„ or the sample rate f, = 1/t, then the frequency axis variable is rn / Nt, = tuf,/ N. So each X(m) sample is associated with a cyclic frequency of mf,/N Hz. . In this case, the resolution of X(m) is UN. The DFT repetition period, or periodicity, is f, Hz, as shown in FIG. --34(b). If we substitute the cyclic frequency variable mf,/N for the generic variable of m/N in (eqn. 47), we obtain an expression the DFT of a symmetrical rectangular function, where K < N, in terms of the sample rate f, in Hz. That expression is: (eqn. 51) For an all ones rectangular function where K = N, the amplitude normalized sin(x)/x approximation in (eqn. 50) can be rewritten in terms of sample rate Is in Hz as (eqn. 52) 13.4.2 DFT Frequency Axis in Radians/Second. We can measure X(m)'s frequency in radians/s by defining the time-domain sample rate in radians/s as co, = 27rf8 . So each X(m) sample is associated with a radian frequency of nicosIN = 2nmfs /N radians/s. In this case, X(m)'s resolution is cos /N = 27cfs /N radian ' s/s, and the DFT repetition period is cos = 27-cf s radians/s, as shown by the expressions in parentheses in FIG. 34(1)). With 6.)5 = 27c4, then rcf s = (.1),/2. If we substitute (.1)5 /2 for nil, in (eqn. 51), we obtain an expression for the DFT of a symmetrical rectangular function, where K <N, in terms of the sample rate cos in (eqn. 53) For an all ones rectangular function where K = N, the amplitude normalized sin(x)/x approximation in (eqn. 50) can be stated in terms of a sample rate cos in radians/s as 2 sin(mcos / 2) X(mo.),) --, mws. (eqn. 54) 13.4.3 DFT Frequency Axis Using a Normalized Angle Variable. Many authors simplify the notation of their equations by using a normalized variable for the radian frequency cos = 2icf s . By normalized, we mean that the sample rate fs is assumed to be equal to 1, and this establishes a normalized radian frequency co s equal to 2n. Thus, the frequency axis of X(m) now becomes a normalized angle co, and each X(m) sample is associated with an angle of mco/N radians. Using this convention, X(m)'s resolution is co/N radians, and the DFT repetition period is co = 2rc radians, as shown by the expressions in brackets in FIG. 34(6). Unfortunately the usage of these three different representations of the DFT's frequency axis is sometimes confusing for the beginner. When reviewing the literature, the reader can learn to convert between these frequency axis notations by consulting FIG. 34 and Table 1. 13.5 Alternate Form of the DFT of an All Ones Rectangular Function Using the normalized radian angle notation for the DFT axis from the bottom row of Table 1 leads to another prevalent form of the DFT of the all ones rectangular function in FIG. 31. Letting our normalized discrete frequency axis variable be o)„, = 2n/n/N, then nm = Nco„,/2. Substituting the term Ne,,,/2 for nut in (eqn. 48), we obtain All Ones form of the Dirichlet kernel (Type 4): (eqn. 55) Equation (eqn. 55), taking the third form of (eqn. 34) sometimes seen in the literature, also has the DFT magnitude shown in FIG. 32(b) and FIG. 32(c). We've covered so many different expressions for the [)FT' of various rectangular functions that it's reasonable to compile their various forms in Table 2. Table 2 DFT of Various Rectangular Functions FIG. 35 General rectangular frequency-domain function of width K samples defined over N samples where K < N. 13.6 Inverse DFT of a General Rectangular Function Let's think now about computing the inverse DFT of a rectangular frequency domain function; that is, given a rectangular X(m) function, find a time domain function x(n). We can define the general form of a rectangular frequency-domain function, as we did for FIG. 24, to be that shown in FIG. 35. The inverse DFT of the rectangular function X(m) in FIG. 35 is (eqn. 56) The same algebraic acrobatics we used to arrive at (eqn. 43) can be applied to (eqn. 56), guiding us to … (eqn. 57) for the inverse DFT of the rectangular function in FIG. 35. The descriptions we gave for (eqn. 43) apply equally well to (eqn. 57) with the exception of a 1/N scale factor term and a change in the sign of (eqn. 43)'s phase angle. Let's use (eqn. 57) to take a 64-point inverse DFT of the 64-sample rectangular function shown in FIG. 36(a). The inverse DFT of the sequence in FIG. 36(a) results in an x(n) whose real and imaginary parts, xreal (n) and Ximig(fl)# are shown in FIG. 36(b) and FIG. 36(c), respectively. With N = 64 and K = 11 in this example, we've made the inverse DFT functions in FIG. 36 easy to compare with those finward DFT functions in FIG. 26. See the similarity between the real parts, X 11 (ni) and xrcai (n), in FIG. 26(b) and FIG. 36(b)? Also notice the sign change between the imaginary parts in FIG. 26(c) and FIG. 36(c). FIG. 36 Inverse DFT of a general rectangular function: (a) original function X(m); (b) real part, xfed(n); (c) imaginary part, ximna(n). FIG. 37 Inverse DFT of a generalized rectangular function: (a) magnitude I x(n) I; (b) phase angle of x(n) in radians. The magnitude and phase angle of x(n) are shown in FIG. 37(a) and FIG. 37(b). Note the differences in main lobe peak magnitude between FIG. 27(a) and FIG. 37(a). The peak magnitude value in FIG. 37(a) is K/N 11/64, or 0.172. Also notice the sign change between the phase angles in FIG. 27(b) and FIG. 37(b). The illustrations in Figures 3-26, 3-27, 3-36, and 3-37 are good examples of the fundamental duality relationships between the forward and inverse DFf. FIG. 38 Frequency-domain rectangular function of width K samples defined over N samples. FIG. 39 Inverse DFT of a rectangular function centered about m = 0: (a) original H(m): (b) hred(n); (c) himag(n); (d) magnitude of h(n); (e) phase of h(n) in radians. 13. 7 Inverse DFT of a Symmetrical Rectangular Function The inverse DFT of the general rectangular function in FIG. 36 is not very common in digital signal processing. However, in discussions concerning digital filters, we will encounter the inverse DFT of symmetrical rectangular functions. This inverse DFT process is found in the window design method of what are called low-pass finite impulse response (FIR) digital filters. That method begins with the definition of a symmetrical frequency function, H(m), such as that in FIG. 38. The inverse DFT of the frequency samples in FIG. 38 is then taken to determine the time domain coefficients for use in a low-pass FIR filter. (Time-domain FIR filter coefficients are typically denoted h(n) instead of x(n), so we'll use h(n) throughout the remainder of this inverse DFT discussion.) In the case of the frequency domain H(m) in FIG. 38, the K unity valued samples begin at m = -m0 = -{K-1)/2. So plugging (K-1)/2 in for m0 in (eqn. 57) gives (eqn. 58) Again, because & 0 = 1, (eqn. 58) becomes (eqn. 59) Equation (eqn. 59) tells us that the inverse DFT of the symmetrical rectangular function in FIG. 38 is itself a real function that we can illustrate by example. We'll perform a 64-point inverse DFT of the sequence in FIG. 39(a). Our H(m) is 11 unity-valued samples centered about them= 0 index. Here the inverse DFT results in an h(n) whose real and imaginary parts are shown in FIG. 39(b) and FIG. 39(c), respectively. As (eqn. 47) predicted, hrea1 (n) is nonzero and hima (n) is zero. The magnitude and phase of h(n) are depicted in FIG. 39(df and FIG. 39(e). (Again, we've made the functions in FIG. 39 easy to compare with the forward DFT function in FIG. 29.) It is h(n)'s real part that we're really after here. Those values of hreal(n) are used for the time-domain filter coefficients in the design of FIR low-pass filters that we'll discuss in Section 5.3. FIG. 40 Complex time-domain sequence xc(n) = eJ'lxnk/N having two complete cycles (k = 2) over N samples: (a) real part of xc(n); (b) imaginary part of xc(n). 14 THE DFT FREQUENCY RESPONSE TO A COMPLEX INPUT In this section, we'll determine the frequency response to an N-point DFT when its input is a discrete sequence representing a complex sinusoid ex pressed as x/n). By frequency response we mean the DFT output samples when a complex sinusoidal sequence is applied to the DFT. We begin by depicting an x,(n) input sequence in FIG. 40. This time sequence is of the form (eqn. 60) where k is the number of complete cycles occurring in the N samples. FIG. 40 shows x(n) if we happen to let k = 2. If we denote our DFT output sequence as X,(m), and apply our x(n) input to the DFT expression in (eqn. 2) we have If we let N = K, n p, and ti = --27t(k-m)/N, (eqn. 61) become, Why did we make the substitutions in (eqn. 61) to get (eqn. 62)? Because, happily, we've already solved (eqn. 62) when it was (eqn. 39). That closed form solution was (eqn. 41) that we repeat here as (eqn. 63) Replacing our original variables from (eqn. 61), we have our answer: DFT of a complex sinusoid: -4 (eqn. 64) FIG. 41 N-point DFT frequency magnitude response to a complex sinusoid having integral k cycles in the N-point time sequence x(n) ej2rcnkIN. Just as we used L'Hopital's rule to find the peak value of the Dirichlet kernel in (eqn. 44), we could also evaluate (eqn. 64) to show that the peak of X,,(nt) is N when nt = k. Like the Dirichlet kernel in (eqn. 43), the Viz) in (eqn. 64) is a complex expression where a ratio of sine terms is the amplitude of X(m) and the exponential term is the phase angle of k(m). At this point, we're interested only in the ratio of sines factor in (eqn. 64). Its magnitude is shown in FIG. 41. Notice that, because x(n) is complex, there are no negative frequency components in Xi .(m). Let's think about the shaded curve in FIG. 41 for a moment. That curve is the continuous Fourier transform of the complex x,.(n) and can be thought of as the continuous spectrum of the x(n) sequence. By continuous spectrum we mean a spectrum that's defined at all frequencies, not just at the periodic fs /N analysis frequencies of an N-point DFT. The shape of this spectrum with its main lobe and sidelobes is a direct and un avoidable effect of analyzing any finite-length time sequence, such as our x(n) in FIG. 40. We can conceive of obtaining this continuous spectrum analytically by taking the continuous Fourier transform of our discrete Mil) sequence, a process some authors call the discrete-time Fourier transform (DTFT), but we can't actually calculate the continuous spectrum on a computer. That's because the DTFT is defined only for infinitely long time sequences, and the DTFT's frequency variable is continuous with infinitely fine-grained resolution. What we can do, however, is use the DFT to calculate an approximation of x,(n)' s continuous spectrum. The DFT outputs represented by the dots in FIG. 41 are a discrete sampled version of xc (n)'s continuous spectrum. We could have sampled that continuous spectrum more often, i.e., approximated it more closely, with a larger DFT by appending additional zeros to the original x(n) sequence. We actually did that in FIG. 21. FIG. 41 shows us why, when an input sequence's frequency is exactly at the m = k bin center, the DFT output is zero for all bins except where = k. If our input sequence frequency was k+0.25 cycles in the sample interval, the DFT will sample the continuous spectrum shown in FIG. 42, where all of the DFT output bins would be nonzero. This effect is a demonstration of DFT leakage described in Section 8. Again, just as there are several different expressions for the DFT of a rectangular function that we listed in Table 2, we can express the amplitude response of the DFT to a complex input sinusoid in different ways to arrive at Table 3. FIG. 42 N-point DFT frequency magnitude response showing spectral leakage of a complex sinusoid having k+0.25 cycles in the N-point time sequence xc(n). Table 3 Various Forms of the Amplitude Response of the DFT to a Complex Input Sinusoid Having k Cycles in the Sample Interval At this point, the thoughtful reader may notice that the DFT's response to a complex input of k cycles per sample interval in FIG. 41 looks suspiciously like the DFT's response to an all ones rectangular function in FIG. 32(c). The reason the shapes of those two response curves look the same is because their shapes are the same. If our DFT input sequence was a complex sinusoid of k = 0 cycles, i.e., a sequence of identical constant values, the ratio of sines term in (eqn. 64) becomes … which is identical to the all ones form of the Dirichlet kernel in (eqn. 48). The shape of our X(m) DFT response is the sinc function of the Dirichlet kernel. 15 THE DFT FREQUENCY RESPONSE TO A REAL COSINE INPUT Now that we know what the DFT frequency response is to a complex sinusoidal input, it's easy to determine the DFT frequency response to a real input sequence. Say we want the DFT's response to a real cosine sequence, like that shown in FIG. 40(a), expressed as (eqn. 71) where k is the integral number of complete cycles occurring in the N samples. Remembering Euler's relationship cos(e) = (ei° + e-i°)/2, we can show the de sired DFT as X(m) where (eqn. 72) Fortunately, in the previous section we just finished determining the closed form of a summation expression like those in (eqn. 72), so we can write the closed form for X(m) as DFT of a real cosine: (eqn. 73) We show the magnitude of those two ratio of sines terms as the sinc functions in FIG. 43. Here again, the DFT is sampling the input cosine sequence's continuous spectrum and, because k = ni, only one DFT bin is nonzero. Because the DFT's input sequence is real, X(m) has both positive and negative frequency components. The positive frequency portion of FIG. 43 corresponds to the first ratio of sines term in (eqn. 73) and the second ratio of sines term in (eqn. 73) produces the negative frequency components of X r(•). DFT leakage is again demonstrated if our input sequence frequency were shifted from the center of the kth bin to k+0.25 as shown in FIG. 44. (We used this concept of real input DFT amplitude response to introduce the effects of DFT leakage in Section 8.) In Table 4, the various mathematical expressions for the (positive frequency) amplitude response of the DFT to a real cosine input sequence are simply those expressions in Table 3 reduced in amplitude by a factor of 2. FIG. 43 N-point DFT frequency magnitude response to a real cosine haying integral k cycles in the N-point time sequence xr (n) = cos(ank/N). FIG. 44 N-point DFT frequency magnitude response showing spectral leak age of a real cosine having k+0.25 cycles in the N-point time sequence xr (n). Table 4 Various Forms of the Positive Frequency Amplitude Response of the DFT to a Real Cosine Input Having k Cycles in the Sample Interval 16. Now that we understand the DFT's overall N-point (or N-bin) frequency response to a real cosine of k cycles per sample interval, we conclude this section by determining the frequency response of a single DFT bias think of a single DFT bin as a kind of bandpass filter, and this useful notion is used, for example, to describe DFT scalloping loss (Section 10), employed in the design of frequency-domain filter banks, and applied in a telephone frequency multiplexing technique known as trans-multiplexing [15]. To determine a DFT single-bin's frequency response, consider applying a real x(n) cosine sequence to a DFT and monitoring the output magnitude of just the m = k bin. Let's say the initial frequency of the input cosine sequence starts at a frequency of k<m cycles and increases up to a frequency of k>m cycles in the sample interval. If we measure the DFT's m = k bin during that frequency sweep, we'll see that its output magnitude must track the input cosine sequence's continuous spectrum, shown as the shaded curves in FIG. 45. FIG. 45(a) shows the m = k bin's output when the input xr(n)'s frequency is k = m-2.5 cycles per sample interval. Increasing xr (n)'s frequency to k = m-1.5 cycles per sample interval results in the ni = k bin's output, shown in FIG. 45(b). Continuing to sweep xr(n)'s frequency higher, FIG. 45(c) shows the ni = k bin's output when the input frequency is k = Throughout our input frequency sweeping exercise, we can see that the ni = k bin's output magnitude must trace out the cosine sequence's continuous spectrum, shown by the solid curve in FIG. 45(d). This means that a DFT's single-bin frequency magnitude response, to a real input sinusoid, is that solid sinc function curve defined by Eqs. (eqn. 74) through (eqn. 79). 17. Now that we've learned about the DFT, it's appropriate to ensure we under stand what the DFT actually represents, and avoid a common misconception regarding its behavior. In the literature of DSP you'll encounter the topics of continuous Fourier transform, Fourier series, discrete-time Fourier transform, discrete Fourier transform, and periodic spectra. It takes effort to keep all those notions clear in your mind, especially when you read or hear someone say something like "the DFT assumes its input sequence is periodic in time." (You wonder how this can be true because it's easy to take the DFT of an aperiodic time sequence.) That remark is misleading at best because DFTs don't make assumptions. What follows is how I keep the time and frequency periodicity nature of discrete sequences straight in my mind. Consider an infinite-length continuous time signal containing a single finite-width pulse shown in FIG. 46(a). The magnitude of its continuous Fourier transform (CFT) is the continuous frequency-domain function )(Jct.)). If the single pulse can be described algebraically (with an equation), then the CFT function Xi (to), also an equation, can be found using Fourier transform calculus. (Chances are very good you actually did this as a homework, or test, problem.) The continuous frequency variable co is radians per second. If the CFT was performed on the infinite-length signal of periodic pulses in FIG. 46(b), the result would be the line spectra known as the Fourier series X2(()). Those spectral lines (impulses) are infinitely narrow and X2 ((o) is well defined in between those lines, because X2 (o) is continuous. (A well known example of this concept is the CFT of a continuous squarewave, which yields a Fourier series whose frequencies are all the odd-multiples of the square wave's fundamental frequency.) FIG. 46(b) is an example of a continuous periodic function (in time) having a spectrum that's a series of discrete spectral components. You're welcome to think of the X2(m) Fourier series as a sampled version of the continuous spectrum in FIG. 46(a). The time-frequency relationship between x2 (t) and X2 (co) shows how a periodic function in one domain (time) leads to a function in the other domain (frequency) that is discrete in nature. Next, consider the infinite-length discrete time sequence x(n), containing several non-zero samples, in FIG. 46(c). We can perform a CFT of x(n) de scribing its spectrum as a continuous frequency-domain function X3(0)). This continuous spectrum is called a discrete-time Fourier transform (DTFT) defined by ] see p. 48 of Reference 16] (eqn. 80) To illustrate the notion of the DTFT, let's say we had a time sequence de fined as x(n) = (0.75)" for n 0. Its DTFT would be (eqn. 81) Equation (eqn. 81) is a geometric series (see Section B) and can be evaluated as (eqn. 82) X0 (w) is continuous and periodic with a period of 2, whose magnitude shown in FIG. 47. This is an example of a sampled (or discrete) time-do main sequence having a periodic spectrum. For the curious reader, we can verify the 2n periodicity of the DTFT using an integer k in the following:
(eqn. 83) because rizmkn = 1 for integer values of k. X3 (w) in FIG. 46(c) also has a 2n periodicity represented by co, = 1-cf 5 , where the cyclic frequency fs is the reciprocal of the time period between the x(n) samples. The continuous periodic spectral function X3 (w) is what we'd like to be able to compute in our world of DSP, but we can't. We're using computers and, sadly, we can't perform continuous signal analysis with the discrete (binary number) nature of computers. All of our processing comprises discrete numbers stored in our computer's memory and, as such, all of our time-domain signals and all of our frequency-domain spectra are discrete sampled sequences. Consequently the CFT, or inverse CFT, of the sequences with which we work will all be periodic. The transforms indicated in FIG. 46(a) through (c) are pencil-and paper mathematics of calculus. In a computer, using only finite-length discrete sequences, we can only approximate the CFT (the DTFT) of the infinite-length x(n) time sequence in FIG. 46(c). That approximation is called the discrete Fourier transform (DFT), and it's the only DSP Fourier tool we have available to us. Taking the DFT of x1 (n), where xl (n) is a finite-length portion of x(n), we obtain the discrete periodic Xi (m) spectral samples in FIG. 46(d). Notice how Xi (m) is a sampled version of the continuous periodic X3 (co). However, Xi (m) is exactly equal to the CFT of the periodic time sequence x2 (n) in FIG. 46(d). So when someone says "the DFT assumes its input sequence is periodic in time" what they really mean is the DFT is equal to the continuous Fourier transform (called the DTFT) of a periodic time domain discrete sequence. After all this rigamarole, the end of the story is this: if a function is periodic, its forward/inverse DTFT will be discrete; if a function is discrete, its forward/inverse DTFT will be periodic.
[1 ] Bracewell, R. "The Fourier Transform," Scientific American, June 1989. [2] Struik, D. A Concise History of Mathematics, Dover Publications Inc., New York, 1967, p. 142. [3] Williams, C. S. Designing Digital Filters. Section 8.6, Prentice-Hall, Englewood Cliffs, New Jersey, 1986, p. 122. [4 ] Press, W., et al. Numerical Recipes-The Art (1 Scientific Computing. Cambridge University Press, 1989, p. 426. [5] Geckinli, N. C., and Yavuz, D. "Some Novel Windows and a Concise Tutorial Comparison of Window Families," IL1.1: Trans. on Acoust. Speech, and Signal Proc., Vol. ASSP-26, No. 6, December 1978. (By the way, on page 505 of this paper, the phrase "such that W(f) 0 V f" indicates that W(f) is never negative. The symbol V means "for all.") [6] O'Donnell, J. "Looking Through the Right Window Improves Spectral Analysis," LPN, November 1984. [7] Kaiser, J. F. "Digital Filters," in System Analysis by Digital Computer. Ed. by F. F. Kuo and J. F. Kaiser, John Wiley and Sons, New York, 1966, pp. 218-277. [8] Rabiner, L. R., and Gold, B. The Theory and Application of Digital Signal Processing. Prentice-I tall, Englewood Cliffs, New Jersey, 1975, p. 88. [9] Schoenwald, J. "The Surface Acoustic Wave Filter: Window Functions," RE Design, March 1986. [10] Harris, E "On the Use of Windows for Harmonic Analysis with the Discrete Fourier Trans form," Proceedings of the IEEE, Vol. 66, No. 1, January 1978. [11] Nuttali, A. H. "Some Windows with Very Good Sidelobe Behavior," IEEE Trans. on & oust. Speech, and Signal Proc., Vol. ASSP-29, No. 1, February 1981. [12] Yanagimoto, Y. "Receiver Design for a Combined RI' Network and Spectrum Analyzer," Hewlett-Packard Journal, October, 1993. [13] Gullemin, E. A. The Mathematics of Circuit Analysis. John Wiley and Sons, New York, 1949, p.511. [14] Lanczos, C. Discourse on Fourier Series, Section 1, Hafner Publishing Co., New York, 1966, p. 7-47. [15] Freeny, S. "TDM/FDM Translation As an Application of Digital Signal Processing," IEEE Communications Magazine, January 1980. [16] Oppenheim, A., et al., Discrete-Time Signal Processing, 2nd ed. Prentice-I kill, Upper Saddle River, New Jersey, 1999, pp. 48-51. PREV. | NEXT Related Articles -- Top of Page -- Home |

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